In the age where the communication market is gradually developing, applications of related IC are growing continuously. Together with the development of portable products such as cell-phones, the utility duration for a battery becomes increasingly important. However, to improve the efficiency while maintaining the stability of a battery is challenging topics. Due to the improvement of the converting efficiency and characteristics of small volume and low noise, low dropout linear regulators (LDO) have become the main stream of low-power dropout and regulation circuits in recent years. They are largely used for communication-related electronic products and various portable systems that are powered by batteries.
Among the existing products (methods), the three-stage operation amplifiers in series are commonly adopted to increase its gain for the purpose of improving the accuracy of low dropout linear regulators, however, such adoption always result in instability. Therefore, various frequency-compensation methods are proposed to increase the systems stability. In the beginning, a large capacitance is used for lowering the position of the dominant pole and to increase the phase margin. However, such circuit has the following drawbacks:
Since the main pole of this circuit falls on the output point, a larger load-capacitance is required to stabilize the system. Nevertheless, it is not easy to integrate the said capacitor into a chip, which increases the difficulty of the system integration.
Generally speaking, a larger system gain is used to promote the system accuracy, but the gain promotion degrades the system stability at the same time. Consequently, either accuracy or stability will have to be sacrificed.
The magnitude of the output current is restricted by the system stability. The larger the output current is, the smaller the load-resistance is, and therefore the dominant pole located at the output point becomes larger correspondingly, which degrades the system stability.
The corresponding first non-dominant pole is lower due to the non-dominant pole of this circuit locates at the output point of the operation amplifier (generally, high impedance). Hence, the system bandwidth is narrower and the transient response is worse.
Therefore, for improving the abovementioned drawbacks, various frequency-compensation methods are continuously proposed, such as nest-type Miller compensation, damping ratio ζ control . . . etc. However, each of these methods needs two compensation capacitors, and the used area of the chip is relatively larger than the area used for the simple Miller compensation method. Consequently, the unitary Miller compensation using gain amplification was proposed latterly. Please refer to FIG. 1, although this method successfully solves the abovementioned problem whereas the damping ratio ζ is affected by the output current such that the stabilization speed of the output voltage is slow down.
Please refer to FIG. 2, which is the small-signal model for FIG. 1. In FIG. 2, gm1, gm2, and gmp are the conductance for the first, the second, and the output stage respectively; g01, g02, and g0L are the output-conductance for the first, the second, and the output stage respectively; Cp1 and Cp2 are the input-parasitic capacitances for the second and the output stage respectively. Cout is the load-capacitance; Re is the parasitic resistance for the load-capacitance; Cm1 and Rm are the compensation capacitance and the compensation resistance respectively; Adc is the DC gain of the system; ζ is the damping ratio. According to the small-signal model of FIG. 2, the transfer function of the system is derived as follow:
                              Av          ⁡                      (            s            )                          =                                                            A                dc                            ⁡                              (                                  1                  +                                                            SC                      out                                        ⁢                                          R                      e                                                                      )                                      ⁡                          [                              1                +                                                      SC                    m1                                    ⁡                                      (                                                                  R                        m                                            -                                                                        g                          o2                                                                                                      g                            m2                                                    ⁢                                                      g                            mp                                                                                                                )                                                  -                                                      S                    2                                    ⁢                                                                                    C                        m1                                            ⁢                                              C                        g                                                                                                            g                        m2                                            ⁢                                              g                        mp                                                                                                        ]                                                          (                              1                +                                  S                                      P                                                                  -                        3                                            ⁢                      db                                                                                  )                        ⁢                          (                              1                +                                                      SC                    out                                    ⁡                                      (                                                                  R                        e                                            +                                                                        g                          o2                                                                                                      g                            m2                                                    ⁢                                                      g                            mp                                                                                                                )                                                  +                                                      S                    2                                    ⁢                                                                                    C                        g                                            ⁢                                              C                        out                                                                                                            g                        m2                                            ⁢                                              g                        mp                                                                                                        )                                                          (        1        )            where
            A      dc        =                            g          m1                ⁢                  g          m2                ⁢                  g          mp                                      g          o1                ⁢                  g          o2                ⁢                  g          L                      ,            P                        -          3                ⁢        db              =                                        C            m1                    ⁢                      g            m2                    ⁢                      g            mp                                                g            o1                    ⁢                      g            o2                    ⁢                      g            L                              .      
                    ζ        =                              1            2                    ⁢                                    C              out                        ⁡                          (                                                R                  e                                +                                                      g                    o2                                                                              g                      m2                                        ⁢                                          g                      mp                                                                                  )                                ⁢                                                                      g                  m2                                ⁢                                  g                  mp                                                                              C                  g                                ⁢                                  C                  out                                                                                                                                  →                      for            ⁢                                                  ⁢            small            ⁢                                                  ⁢            ESR                          ⁢                  ≈                                    1              2                        ⁢                          g              o2                        ⁢                                                            C                  out                                                                      g                    m2                                    ⁢                                      g                    mp                                    ⁢                                      C                    g                                                                                                          (        2        )                                α        ⁢                                            I              b2                                      g              mp                                                          (        3        )                                (                              gm2            =                                          I                b2                                                              V                  gs2                                -                                  V                  th2                                                              ,                      go2            =                          λ              ⁢                                                          ⁢              Ib2                                      )                                        
For reducing the cost, the recent market tends to use cheaper capacitors such as ceramic capacitors as the load-capacitors. Because the parasitic resistance of a ceramic capacitor is smaller, so Eq. (2) can be simplified to Eq. (3).
Knowing from Eq. (3), the damping ratio ζ is inverse-proportional to the conductance of the output stage (gmp), and is proportional to the bias current of the 2nd stage amplifier (Ib2). Since the conductance of the output stage (gmp) getting larger (approximately 30 times) if the output current getting larger (eg. from 0.1 mA to 150 mA), the damping ratio ζ decrease to become smaller than 1 (or even far smaller than 1). Accordingly, the frequency response has a surge around the unit-gain frequency, which leads to a ripple on the transient response of the output voltage Vout when the output current suddenly changes such that the stabilization speed of the output voltage is slow down. As a result, it cannot provides a quick-recovery low dropout linear regulator.